Specialty 221600
Saint Petersburg
1. PURPOSE OF THE WORK
The purpose of this work is to study the principle of operation and determine the effectiveness of the suppressor of pulsed broad-spectrum interference.
2. BRIEF INFORMATION FROM THE THEORY
The main methods for protecting radio receivers from pulsed broad-spectrum interference are:
a) non-receiving - the use of narrowly directed antennas, removing the antenna from the zone of impulse interference and suppressing interference at the place of their occurrence;
b) circuit - various ways of processing a mixture of useful signal - impulse noise in order to weaken the interfering effect.
One of the effective circuit methods of dealing with impulse noise is the use of a wide band - amplitude limiter - narrow band scheme (SHOU scheme). Such a scheme is often used in radio communications.
In this paper, we study the SHOW scheme for two cases:
a) the useful signal is video pulses;
b) the useful signal is a continuous radio signal with amplitude modulation.
Structural diagrams for these cases are shown in fig. 1a and 1b, respectively. In the first case, the SHOU circuit is located after the BP amplitude detector, in the second case, in the radio frequency path to the BP.
The SHOW scheme presented in fig. 1a includes a wideband video amplifier, an amplitude limiter and a narrowband video amplifier connected in series. At the input of the circuit: a signal-interference mixture comes from the detector (Fig.2a), and the signal duration is much greater than the interference duration (tc>>tp), and the interference amplitude is much greater than the signal amplitude (Up>>Uc). The broadband amplifier is designed to amplify the input mixture to a level that ensures the normal operation of the limiter. The bandwidth of the amplifying path to the limiter is chosen so as to avoid a significant increase in the duration of the interference pulse (Fig. 2b). The clipping threshold is slightly higher than the useful signal level; therefore, after clipping, the signal and noise levels become almost equal (Fig. 2c). A narrow-band video amplifier (or filter) acts as an integrator whose time constant is matched to the signal duration and far exceeds the noise duration. Due to the fact that tc>>tp, the signal at the filter output has time to grow to its amplitude value, but the noise does not (Fig. 2d). Thus, the signal-to-noise ratio at the output of the SHOW circuit increases dramatically.
Let us estimate the gain in the signal/noise ratio when using the SHOW scheme. At the input of the circuit, there is a signal with amplitude Uc and duration tc and interference with a rectangular envelope (Up, tp). The role of the integrator is performed by an RC - first-order circuit with a transient response of the form
h(t)=1- exp(- tP/ tRC) (1)
where tRC = RC is the filter time constant.
It is known from theory that the duration of the signal rise to the level of 0.9 Uc for such a circuit is determined by the relation
t n=2.3 t RC (2)
The noise level at the output of the amplitude limiter Uп = Ulimit, where Ulimit is the limitation threshold, and the level of the useful signal and noise at the output of the circuit, respectively
Ucexit=0,9 uck (3)
Upout= UogreK (4)
where K is the gain of the circuit. Voltage signal-to-noise ratio at the output of the SHOW circuit
hexit=(Uc/ UP)out=0.9*Ufrom/(Uogre) (5)
The gain from using the scheme is determined by the relation
(6)
or, taking into account (5),
q1 =0.9* UP/(Uogre(1/)) (7)
Because tP<< tRC Andtfrom=2,3 tRC, then
q1 =(0.9* UP/ Uogre)*(tfrom/2,3 tP) » 0.4( UP/ Uogre)*(tfrom/ tP) (8)
With the SHOW circuit turned off (limiter off), the noise level at the output
Upout= UPK (9)
In this case, the signal-to-noise ratio at the output
hexit=(Uc/ UP)out=0.9*Ufrom/(UP) (10)
and the gain obtained due to the "narrow band" of the output filter, matched in band with the useful signal, is equal to
q2=[ hexit/ hin]SHOWoff=0.9/ (11)
The relative gain obtained when using the SHOW scheme is defined as the ratio
n= q1/ q2 (12)
After substituting (7) and (11) into (12) and taking into account the relations
n<< tRC Andtfrom=2,3 tRC, , we have
n= q1/ q2 = UP/ Uogre (13)
In the SHO scheme (Fig. 16), the broadband amplifier is the resonant stages of the intermediate frequency amplifier (IFA) with a bandwidth much wider than the useful signal spectrum width. The IF is located up to the limiter. An IF cascade after the limiter is used as an integrator, and the bandwidth of this cascade is matched to the width of the spectrum of the useful signal. To avoid deterioration of the receiver noise immunity due to the expansion of the bandwidth of the IF stages to the limiter, the SHOU circuit is located as close as possible to the receiver input.
3. DESCRIPTION OF THE LABORATORY SETUP
The block diagram of the laboratory setup for studying the noise suppressor is shown in fig. 3. The composition of the laboratory installation includes:
1. Generator of standard signals (GSS);
2. Oscilloscope;
3. Laboratory model of an interference suppressor.
The block diagram of the installation is shown in fig. 4. The circuit contains a simulator of a mixture of signals and noise and a SHOW circuit. An amplitude-modulated oscillation (AMW) from the GSS is fed to the input of the simulator of the mixture of signal and impulse noise. AMK has the following parameters:
a) amplitude Um = 100 mV;
b) carrier frequency fo == 100 kHz;
c) modulation frequency fm = 1 kHz. The simulator generates the following signals:
Sam - useful AMK;
Si - pulse useful signal;
Sp - rectangular impulse noise;
Spp - radio pulse interference with a rectangular shape of the envelope.
SYNC - oscilloscope clock pulse. On the front panel of the laboratory layout, it is possible to switch on the simulated signals and noises by using the "Signal on" and "Noise on" toggle switches, respectively. The useful impulse signal is mixed with the impulse noise in the adder å1, and the continuous useful signal from AM and radio impulse noise - in the adder å2. A mixture of a useful signal with interference is fed to two SHOW circuits designed to operate both at video frequency and at radio frequency. Switching circuits is carried out by the "Sam-Si" switch located on the front panel of the layout. The first circuit contains a broadband video amplifier (SHVU), a limiter, based on diodes VD1, VD2, and a narrow-band filter (UV1) implemented by an RC circuit. The second circuit contains a broadband amplifier, a limiter, a narrow band filter (UV2), and an AMK detector. UV2 is an oscillatory circuit L1 Sk1 Sk2, the bandwidth of which is matched with
the width of the AMC spectrum. The limiter is turned on by the toggle switch "ON PP". The three-position test point switch (1, 2, 3) allows you to use an oscilloscope to observe the signals at the input of the SHOW circuit, at the input of the limiter and at the output of the circuit.
4. ORDER OF PERFORMANCE OF WORK
3.1. Familiarize yourself with the principle of operation of the interference suppressor and the composition of the equipment used.
3.2. Investigation of an interference suppressor in the presence of a pulsed useful signal.
3.2.1. Preparation for work:
Set a signal at the GSS output with the following parameters:
a) amplitude - 100 mV;
b) frequency - 100 kHz;
c) modulation depth - 30%.
Turn on the layout, set the "Sam-Si" switch to the Si position, the "Interference on", "Signal on" switches - to the on position, the control point switch - to position 1.
3.2.2. Measurements:
Using an oscilloscope, measure the parameters of the signal and noise at the input of the circuit (signal amplitude Uc and noise Upp; signal duration tc and noise tp);
Calculate the signal-to-noise ratio from the voltage at the input of the circuit;
Observe the signal at the control points of the circuit with the noise suppressor turned on and off, turning off the limiter with the "On PP" toggle switch;
Measure the signal-to-noise ratio at the output of the circuit with the noise suppressor on and off;
Based on the measurement results, determine the relative gain and compare with the calculated one;
Draw oscillograms at the control points of the circuit with the suppressor on and off.
3.3.Research of the interference suppressor when receiving a continuous cAM signal.
3.3.1. Preparation for work:
Set the switches to the following positions:
a) "Sam-Si"-Sam
b) "Signal on" - enabled;
c) "Interference on" - off;
d) control points - 3;
by changing the generator frequency within 100 kHz, to achieve the maximum signal at the output of the detector. Observation is carried out on the oscilloscope screen.
3.3.2 Measurements:
Observe the signal at the control points of the circuit with the noise suppressor turned on and off, turning off the limiter with the "On PP" toggle switch,
Measure the signal-to-noise ratio at the input of the circuit (test point 1);
Measure the signal-to-noise ratio at the output of the circuit (test point 3) with the suppressor on and off;
Note, the levels of the useful signal and noise at the input and output of the circuit are measured separately (the signal and noise are switched on by the "signal on" and "noise on" toggle switches);
Based on the measurement results, determine the gain in relation to the signal noise when using the SHOW scheme and the relative gain.
block diagram of the studied noise suppressor;
oscillograms of signals at the control points of the circuit;
calculation of the expected gain in terms of signal/interference when receiving video signals;
experimental data on the effectiveness of the interference suppressor for video and radio signals.
LITERATURE
Protection against radio interference. , and etc.; Ed. M.: Sov. radio, 1976
Surge suppressor for Р399А.
Over the past few months, with the inclusion of street lighting, it has become almost impossible for me to work on the air due to the presence of strong interference from DRL lamps. My device is not imported, but a transceiver R399A, which is used as a base unit for VHF (“Hyacinth” is used as a reference oscillator in HF synthesizers for consoles). Having gone on vacation, I decided to somehow deal with the problem that had arisen, and within a week the proposed “Suppressor of impulse noise (PIP)” was designed.
Schematic diagram of the device is shown in Fig.1. The PIP consists of two nodes: a peak detector and a pulse suppression node. The device between the second mixer and the IF is switched on (215 kHz path).
The peak detector circuit with some modifications was borrowed from the magazine “Ham Radio, 2, 1973, W2EGH”, in particular, the chains D1, R6, S1 and D2, R7, S2 were added, and the suppressor assembly was made according to the controlled attenuator circuit R16, C18, Q4, the introduction of which, among other things, somewhat improved the dynamic range of the receiver's AGC. The use of delay lines common for these LC devices did not give any advantage. Probably due to their narrow bandwidth due to the low IF and, as a result, the “stretching” of the interference pulse. The use of a broadband amplifier based on the KT610A transistor at the input of the peak detector is due to the need to obtain an undistorted signal at the output with an amplitude of up to 20 V and, accordingly, a minimal effect on the duration and shape of the initial noise pulse. The use of additional AGC in the amplifier only worsened its operation, but the introduction of the D2, R7 chain automatically blocks the operation of the PIP in the presence of a powerful useful signal (tested up to +60 dB on a real signal from the air with full amplification of R1). S1 - “Deep suppression” allows you to eliminate even small interference only at very low levels of the useful signal (tested when receiving EME stations in the JT65B mode), with a signal strength of S2 or more, the detected envelope overlaps the signal. The quality of decoding in the FSK441 mode has not really been tested yet.
The PIP scheme is still in the process of being finalized, but, nevertheless, it can already provide a good service for real work on the air to those who need it. Any revision and publication that improves the parameters of the device is also welcome.
In switching power supplies, interference occurs when switching key elements. This interference is induced on the power cable connected to the AC mains. Therefore, measures must be taken to suppress them.
Typical EMI Mains Filter Solution for Switching Power Supply
To suppress interference penetrating through the power cable into the primary circuit from a switching power supply, the circuit shown in Figure 9 is used.
Figure 9 - Suppression of noise penetrating through the cable
Differential and common mode noise
There are two types of interference: differential and common mode. The differential noise current induced on both wires of the power line flows in opposite directions in them, as shown in Figure 10. The common mode current flows in all lines in the same direction, see Figure 11.
Figure 10 - Differential noise
Figure 11 - Common mode noise
Functional purpose of network filter elements
The figures below show examples of the use of various filter elements and graphics illustrating the effect of their application. The graphs shown show the change in the intensity of differential and common-mode noise of a switching power supply relative to the level of industrial noise. Figure 12 shows the signal graphs in the absence of a filter at the input of a switching power supply. As can be seen from the graph, the level of differential and common mode noise is quite high. Figure 13 illustrates an example of using a filtering X-capacitor. The graph shows a noticeable decrease in the level of differential noise.
Figure 14 shows the results of using X-capacitors and Y-capacitors together. The graph clearly shows the effective suppression of both common mode and differential noise. The use of X-capacitors and Y-capacitors in combination with a common mode choke (common mode choke) is shown in Figure 15. The graph shows a further reduction in the level of both differential and common mode noise. This is because a real common mode choke has some differential inductance.
Figure 12 - Without filter
Figure 13 - Using an X-capacitor
Figure 14 - Using an X-capacitor and a Y-capacitor
Figure 15 - Using an X-capacitor, a Y-capacitor and a common mode choke
An example of interference suppression in a mobile phone
Sources of radiated interference
The interference generated by the signal processing unit passes into the RF unit, which leads to a significant deterioration in sensitivity. A mobile phone signal processing unit, which is usually built on a baseband signal processing IC, controls various signals such as a voice signal and a signal for an LCD display. A signal processing IC is a source of significant interference because it operates at a high frequency and has many data lines connected to it. When interference passes through the data lines or power/GND buses from the signal processing unit to the RF unit, its sensitivity deteriorates, as a result, the Bit Error Rate (BER) increases.
Components for interference suppression in mobile phones
To improve the BER parameter (Bit Error Rate), that is, to reduce the percentage of received erroneous bits, it is necessary to suppress interference from the signal processing unit into the RF unit. To do this, install EMI filters on all buses connecting these blocks. In addition, it is also important to shield the signal processing unit, since the level of interference emitted by it has increased significantly in the latest models of mobile phones.
Setting Filters on the Display Control Bus
The LCD control bus contains many signal lines switching at the same time, which causes a significant increase in the surge current flowing in the ground (GND) and power circuits. Therefore, it is necessary to limit the current flowing through the signal lines. Typically, BLA31 series ferrite chip bead arrays and NFA31G series EMIFIL® chip filters with a resistor are used for this. If, for structural reasons, the use of these components is not possible, then EMC absorbers of the EA series should be used to suppress interference passing through the flexible cable of the LCD display.
Shielding improvement
Typically, a conductive coating is applied to the inner surface of the plastic housing of a mobile phone. With the expansion of the functionality of the mobile phone, the level of interference from the signal processing unit also increases. Therefore, it is necessary to shield the signal processing unit with the same care as the RF unit. When designing a mobile phone case, in order to reduce the impedance at high frequency, one should try to ensure that the contact area between the parts of the case is as large as possible. To improve shielding, metal shielding elements or EMC absorbers should be used where possible in the signal processing unit.
Shevkoplyas B.V. «Microprocessor structures. Engineering solutions.» Moscow, Radio publishing house, 1990. Chapter 4
4.1. Interference suppression on the primary supply network
The waveform of the alternating voltage of an industrial power supply (~ "220 V, 50 Hz) for short periods of time can differ greatly from a sinusoidal one - surges or "inserts" are possible, a decrease in the amplitude of one or more half-waves, etc. The causes of such distortions are related usually with a sharp change in the network load, for example, when a powerful electric motor, furnace, welding machine is turned on.Therefore, it is necessary, if possible, to isolate from such sources of interference through the network (Fig. 4.1).
Rice. 4.1 Options for connecting a digital device to the primary power supply
In addition to this measure, it may be necessary to introduce a line filter at the power input of the device in order to suppress short-term interference. The resonant frequency of the filter can be in the range of 0.1.5-300 MHz; broadband filters provide interference suppression over the entire specified range.
Figure 4.2 shows an example of a network filter circuit. This filter has dimensions of 30 X30X20 mm and is mounted directly on the network input block into the device. The filters should use high frequency capacitors and inductors either without cores or with high frequency cores.
In some cases, it is mandatory to introduce an electrostatic shield (an ordinary water pipe connected to a grounded power supply housing) for laying the wires of the primary supply network inside it. As noted in, the short-wave transmitter of the taxi fleet, located on the opposite side of the street, is capable, with a certain relative orientation, of inducing signals with an amplitude of several hundred volts on a piece of wire. The same wire, placed in an electrostatic shield, will be reliably protected from this kind of interference.
Rice. 4.2. Network Filter Circuit Example
Consider methods for suppressing network interference directly in the power supply of the device. If the primary and secondary windings of a power transformer are located on the same coil (Fig. 4.3, a), then due to the capacitive coupling between the windings, impulse noise can pass from the primary circuit to the secondary. According to recommended four ways to suppress such interference (in order of increasing efficiency).
- The primary and secondary windings of the power transformer are carried out on different coils (Fig. 4.3, b). The through capacitance C decreases, but the efficiency decreases, since not all of the magnetic flux from the primary winding region enters the secondary winding region due to scattering through the surrounding space.
- The primary and secondary windings are made on the same coil, but are separated by a copper foil screen with a thickness of at least 0.2 mm. The screen should not be a short-circuited coil. It is connected to the body ground of the device (Fig. 4.3, c)
- The primary winding is completely enclosed in a screen, which is not a short-circuited coil. The screen is grounded (Fig. 4.3, G).
- The primary and secondary windings are enclosed in individual screens, between which a separating screen is laid. The entire transformer is enclosed in a metal case (Fig. 4.3,<Э). Экраны и корпус заземляются. Этот тип трансформатора в силу предельной защищенности от прохождения помех получил название «ультраизолятор».
With all the listed methods of interference suppression, the wiring of network wires inside the device should be carried out with a shielded wire, connecting the shield to the chassis ground. Invalid uk
laying in one bundle of network and other (power boards, signal, etc.) wires "even in the case of shielding of both.
It is recommended to install a capacitor with a capacity of approximately 0.1 μF in parallel with the primary winding of the power transformer in the immediate vicinity of the winding terminals and, in series with it, a current-limiting resistor with a resistance of about 100 ohms. This makes it possible to "close" the energy stored in the core of the power transformer at the moment of opening the mains switch.
Rice. 4.3. Options for protecting the power transformer from the transmission of impulse noise from the network to the secondary circuit (and vice versa):
a - no protection; b - separation of the primary and secondary windings; in- laying the screen between the windings; G - full shielding of the primary winding; e - full shielding of all elements of the transformer
Rice. 4.4. Simplified power supply diagram (but) and diagrams (b, c), explaining the operation of a full-wave rectifier.
The power supply is the greater source of impulse noise over the network, the larger the capacitance of the capacitor C
Note that with an increase in the capacity C of the filter (Fig. 4.4, a) of the power supply of our device, the probability of failures of neighboring devices increases, since the energy consumption from the network by our device increasingly takes on the character of shocks. Indeed, the voltage at the output of the rectifier also increases during those time intervals when energy is taken from the network (Fig. 4.4, b). These intervals in Fig. 4.4 are shaded.
With an increase in the capacitance of the capacitor C, the periods of its charge become smaller (Fig. 4.4, c), and the current taken from the network in a pulse is increasing. Thus, an outwardly “harmless” device can create interference in the network that is “not inferior” to interference from a welding machine.
4.2. Grounding Rules for Protection from Ground Interference
In devices made in the form of structurally finished blocks, there are at least two types of "ground" buses - case and circuit. The case bus, in accordance with safety requirements, must be connected to the ground bus laid in the room. The circuit bus (relative to which the voltage levels of the signals are measured) should not be connected to the case bus inside the block—a separate terminal isolated from the case must be brought out for it.
Rice. 4.5. Incorrect and correct grounding of digital devices. Shown is a ground bus, which is usually available indoors.
On fig. 4.5 shows options for incorrect and correct grounding of a group of devices that are interconnected by information lines. (these lines are not shown). The "ground" circuit buses are connected by individual wires at point A, and the case buses are connected at point B, as close as possible to point A. Point A may not be connected to the ground bus in the premises, but this creates inconvenience, for example, when working with an oscilloscope, which has The "ground" of the probe is connected to the case.
In case of incorrect grounding (see Fig. 4.5), the impulse voltages generated by the equalizing currents on the earth bus will actually be applied to the inputs of the receiving main elements, which can cause their false operation. It should be noted that the choice of the best grounding option depends on the specific "local" conditions and is often carried out after a series of careful experiments. However, the general rule (see Figure 4.5) always holds.
4.3. Interference suppression on secondary power supply circuits
Due to the finite inductance of the power and ground rails, surge currents cause surge voltages of both positive and negative polarity to be applied between the power and ground pins of the ICs. If the power and ground buses are made of thin printed or other conductors, and high-frequency decoupling capacitors are either completely absent, or their number is insufficient, then when several TTL microcircuits are simultaneously switched at the “far” end of the printed circuit board, the amplitude of impulse noise on the power supply (voltage surges acting between the power and ground pins of the microcircuit) can be 2 V or more. Therefore, when designing a printed circuit board, the following recommendations must be followed.
- The power and ground rails must have minimal inductance. To do this, they are made in the form of lattice structures covering the entire area of the printed circuit board. It is unacceptable to connect TTL microcircuits to the bus, which is a "branch", because as it approaches its end, the inductance of the power supply circuits accumulates. The power and ground rails should, if possible, cover the entire free area of the printed circuit board. Particular attention should be paid to the design of dynamic memory storage matrices based on K565RU5, RU7, etc. The matrix should be a square so that the address and control lines have a minimum length. Each microcircuit must be located in an individual cell of a lattice structure formed by power and ground buses (two independent lattices). The power and ground buses of the storage matrix should not be loaded with “foreign” currents flowing from addressable shapers, control signal amplifiers, etc.
- Connecting external power and ground buses to the board through the connector must be made through several contacts evenly spaced along the length of the connector, so that the input to the lattice structures of the power and ground buses is made from several points at once.
- Suppression of power interference should be carried out near the places of their occurrence. Therefore, a high-frequency capacitor with a capacity of at least 0.02 microfarads must be located near the power pins of each TTL chip. This also applies to a particular extent to the mentioned dynamic memory chips. To filter low-frequency interference, it is necessary to use electrolytic capacitors, for example, with a capacity of 100 μF. When using dynamic memory microcircuits, electrolytic capacitors are installed, for example, at the corners of the storage matrix or in another place, but near these microcircuits.
Accordingly, instead of high-frequency capacitors, special power buses BUS-BAR, CAP-BUS are used, which are laid under the lines of microcircuits or between them, without violating the usual automated technology for installing elements on the board, followed by “wave” soldering. These buses are distributed capacitors with a capacitance of approximately 0.02 uF/cm. For the same total capacitance as discrete capacitors, the busbars provide significantly better noise rejection at higher packing densities.
Rice. 4.6. Options for connecting boards P1-PZ to the power supply
On fig. 4.6 provides recommendations for connecting devices made on printed circuit boards P1-PZ to the output of the power supply. A high-current device made on a PZ board creates more noise on the power and ground buses, so it should be physically closer to the power supply, or even better, be powered using individual buses.
4.4. Rules for working with agreed communication lines
On fig. 4.7 shows the shape of the signals transmitted over the cable, depending on the ratio of the resistance of the load resistor R and the wave impedance of the cable p. Signals are transmitted without distortion at R=p. The characteristic impedance of a particular type of coaxial cable is known (eg 50, 75, 100 ohms). The characteristic impedance of flat cables and twisted pairs is usually close to 110-130 ohms; its exact value can be obtained experimentally by selecting a resistor K, when connected, the distortion is minimal (see Fig. 4.7). When conducting an experiment, wire-wound variable resistances should not be used, since they have a large inductance and can introduce distortions into the waveform.
Communication line of the "open collector" type (Fig. 4.8). For transmission of each main signal with a front duration of about 10 ns at distances exceeding 30 cm, a separate twisted pair is used or one pair of cores is allocated in a flat cable. In the passive state, all transmitters are turned off. When any transmitter or group of transmitters is triggered, the voltage on the line drops from a level exceeding 3 V to approximately 0.4 V.
With a line length of 15 m and with its correct matching, the duration of transient processes in it does not exceed 75 ns. The line implements the OR function with respect to signals represented by low voltage levels.
Rice. 4.7. Transmission of signals by cable. O—voltage pulse generator
Communication line of the "open emitter" type (Fig. 4.9 "). This example shows a variant of a line using a flat cable. Signal wires alternate with ground wires. Ideally, each signal wire is flanked on both sides by its own ground wires, but this is usually not necessary. In Figure 4.9, each signal wire is adjacent to "own" and "foreign" ground, which is usually quite acceptable. A flat cable and a set of twisted pairs are essentially the same thing, and yet the latter is preferable in conditions of a high level of external interference. An open-emitter line implements a wire-OR function with respect to signals represented by high voltage levels. The timing characteristics approximately correspond to those of an "open collector" line.
Communication line of the "differential pair" type (Fig. 4.10). The line is used for unidirectional signal transmission and is characterized by increased noise immunity, since the receiver reacts to the signal difference, and the interference induced from the outside acts on both wires in approximately the same way. The line length is practically limited by the ohmic resistance of the wires and can reach several hundred meters.
Fig, 4.8. Communication line of the "open collector" type 
Rice. 4.9. Communication line of the "open emitter" type 
Rice. 4.10. Communication line type "differential pair"
All of the considered lines should use receivers with high input impedance, low input capacitance, and preferably with a hysteresis transfer characteristic to increase noise immunity.
Physical implementation of the highway (Fig. 4. II), Each device connected to the trunk contains two connectors. A scheme similar to that shown in fig. 4.11, was considered earlier (see Fig. 3.3), so we will focus only on the rules that must be observed when designing matching units (SB).
Transmission of trunk signals through connectors. The best options for soldering connectors are shown in fig. .4.12. The front of the pulse running along the main line in these cases almost “does not feel” the connector, since the inhomogeneity introduced into the cable line is insignificant. In this case, however, it is required to occupy 50% of the used contacts underground.
If for some reason this condition is not feasible, then at the expense of noise immunity, it is possible to take the second, more economical, in terms of the number of contacts, option for soldering the connectors, shown in Fig. 4.13. This option is often used in practice. Twisted pair grounds (or flat cable grounds) are assembled on metal strips of the largest possible cross section, for example 5 mm2.
The desoldering of these lands is carried out evenly along the length of the bar, as the corresponding signal wires are desoldered. Both strips are connected through a connector using a series of jumpers of minimum length and maximum cross section, and the jumpers are evenly spaced along the length of the strips. Each ground jumper should correspond to no more than four signal lines, but the total number of jumpers should not be less than three (one in the center and two at the edges).
Rice. 4.13. Permissible option for signal transmission through the connector. H-=5 mm2—bar cross-section, 5^0.5 mm2—ground wire cross-section 
Rice. 4.14. Variants of execution of branches from the main
Implementation of branches from the highway. On fig. 4.14 shows options for incorrect and correct execution of a branch from the main. The path of one line is traced, the earth wire is shown conditionally. The first option (a typical mistake of novice circuit engineers!) is characterized by splitting the wave energy into two parts, 
Rice. 4.15. Options for connecting receivers to the trunk
coming from line A. One part goes to the charge of line B, the other to the charge of line C. After the charge of line C, the “full” wave begins to propagate along line B, trying to catch up with the wave with half the energy that has left earlier. The signal front thus has a stepped shape.
With proper branching, line segments A, C and B are connected in series, so the wave practically does not split and the signal fronts are not distorted. Transmitters and receivers located on the board should be as close as possible to its edge to reduce the inhomogeneity introduced at the merging point of line segments B and C.
One or two-way transceivers can be used to decouple receiver beams from the backbone (see Fig. 3.18. 3.19). When branching the line into several directions, a separate transmitter should be allocated for each (Fig. 4.15, in).
For line transmission, it is better to use not rectangular, but trapezoidal pulses. Signals with shallow fronts, as noted, propagate along the line with less distortion. In principle, in the absence of external interference, for any arbitrarily long and even inconsistent line, one can choose a signal slew rate so slow that the transmitted and received signals will differ by an arbitrarily small amount.
To receive trapezoidal pulses, the transmitter is made in the form of a differential amplifier with an integrating feedback circuit. At the input of the main receiver, also made in the form of a differential amplifier, an integrating circuit is installed to filter high-frequency noise.
When transmitting signals within the board, when the number of receivers is large, "serial matching" is often used. It consists in the fact that in series with the output of the transmitter, in the immediate vicinity of this output, a resistor with a resistance of 20-50 ohms is connected. This makes it possible to suppress oscillatory processes at the signal fronts. This technique is often used when transmitting control signals (KA5, SAZ, \UE) from amplifiers to dynamic memory LSI.
4.5. About the protective properties of cables
On fig. 4.16a shows the simplest scheme for transmitting signals over a coaxial cable, which in some cases can be considered quite satisfactory. Its main disadvantage is that in the presence of pulsed equalizing currents between body grounds (potential equalization is the main function of the body earth system), some of these currents 1 can flow through the cable sheath and cause a voltage drop (mainly due to the inductance of the sheath), which ultimately acts on the load K.
Moreover, in this sense, the circuit shown in Fig. 4.16, a, turns out to be preferable, and with an increase in the number of points of contact between the cable braid and the body ground, the possibilities for induced charges to drain from the braid improve. Using a cable with an additional braid (Fig. 4.16, c) allows you to protect yourself from both capacitive pickups and equalizing currents, which in this case flow through the outer braid and practically do not affect the signal circuit.
The inclusion of a cable with an additional braid according to the scheme shown in fig. 4.16, d, allows you to improve the frequency properties of the line by reducing its linear capacitance. In the ideal case, the potential of any elementary section of the central core coincides with the potential of the elementary cylinder of the inner braid surrounding this section.
Lines of this type are used in local computer networks to increase the speed of information transfer. The outer sheath of the cable is part of the signal circuit, and therefore this circuit is equivalent in terms of protection against external interference to the circuit shown in fig. 4.16.6.
Rice. 4.16. Cable options
Neither the copper nor aluminum sheathing of a simple coaxial cable protects it from exposure to low-frequency magnetic fields. These fields induce EMF both on the segment of the braid and on the corresponding segment of the central core.
Although these EMFs are of the same name in sign, they do not compensate each other in magnitude due to the different geometry of the corresponding conductors - the central core and the braid. The differential EMF is ultimately applied to the load K. An additional braid (Fig. 4. 16, c, d) also unable to prevent the low frequency magnetic field from penetrating into its inner area
Protection against low-frequency magnetic fields is provided by a cable containing a twisted pair of wires enclosed in a braid (Fig. 4.16, e). In this case, the EMF induced by an external magnetic field on the wires that make up the twisted pair completely compensate each other both in sign and in absolute value.
This is all the more true, the smaller the pitch of the wires in comparison with the area of action of the field and the more carefully (symmetrically) the twisting is performed. The disadvantage of such a line is its relatively low frequency "ceiling"—on the order of 15 MHz—because of the large energy losses of the useful signal at higher frequencies.
The scheme shown in fig. 4.16, e, provides the best protection against all types of interference (capacitive interference, equalizing currents, low-frequency magnetic fields, high-frequency electromagnetic fields).
It is recommended to connect the inner braid to the “radio engineering” or “true” (in the literal sense, grounded) earth, and the outer braid to the “system” (circuit or case) earth. In the absence of a "true" ground, you can use the switching circuit shown in fig. 4. 16, well.
The outer braid connects to system ground at both ends, while the inner braid connects to the source side only. In those cases where there is no need to protect against low-frequency magnetic fields and it is possible to transmit information without the use of double-phase signals, one of the twisted pair wires can serve as a signal wire, and the second as a screen. In these cases, the circuits shown in Fig. 4.16, c, g, can be thought of as coaxial cables with three shields - a twisted-pair ground wire, inner and outer cable sheaths.
4.6. Using Optocouplers to Suppress Interference
If the devices of the system are separated by a considerable distance, for example, by 500 m, then it is difficult to count on the fact that their lands always have the same potential. As noted, equalizing currents through earth conductors create impulse noise on these conductors due to their inductance. This interference is ultimately applied to the inputs of the receivers and may cause them to spuriously operate.
The use of "differential pair" type lines (see § 4.4) only suppresses common-mode interference and therefore does not always bark positive results. On fig. 4.17 shows diagrams of optocouplers between two devices remote from each other.
Rice. 4.17. Schemes of optocouplers between devices remote from each other:
a - with an active receiver, b- with active transmitter
The circuit with an "active receiver" (Fig. 4.17, a) contains a transmitting optocoupler VI and a receiving optocoupler V2. When pulse signals are applied to input X, the LED of optocoupler VI periodically emits light, as a result, the output transistor of this optocoupler periodically saturates and the resistance between points a and b drops from several hundred kilo-ohms to several tens of ohms.
When the output transistor of the transmitting optocoupler is turned on, the current from the positive pole of the source U2 passes through the LED of the optocoupler v2, line (points a and b) and returns to the negative pole of this source. Source U2 runs isolated from source U3.
If the output transistor of the transmitting optocoupler is turned off, then no current flows through the source circuit U2. The signal X" at the output of optocoupler V2 is close to zero if its LED is on, and close to +4 V if this LED is off. Thus, at X==0, the LEDs of the transmitting and receiving optocouplers are on and, therefore, X"==0. With X==1, both LEDs are off and X"==1.
Optocoupler isolation can significantly increase the noise immunity of the communication channel and ensure the transmission of information over distances of the order of hundreds of meters. Diodes connected to the transmitting and receiving optocouplers serve to protect them from reverse voltage surges. The resistor circuit connected to the source U2 serves to set the current in the line and limit the current through the LED of the receiving optocoupler.
The current in the line according to the IRPS interface can be selected equal to 20 or 40 mA. When choosing resistor values, the ohmic resistance of the communication line must be taken into account. Scheme with an "active transmitter" (Fig. 4.17, b) differs from the previous one in that the power supply of the U2 line is located on the side of the transmitter. This does not provide any advantages - both circuits are essentially the same and are the so-called "current loops".
The recommendations given in this chapter may seem too harsh to the novice circuit designer. The fight against interference seems to him to be a “fight with a windmill”, and the lack of experience in designing devices of increased complexity creates the illusion that it is possible to create a workable device without following any of the above recommendations.
Indeed, this is sometimes possible. There are even cases of serial production of such devices. However, in informal reviews of their work, you can hear many interesting non-technical expressions, such as visit effect and some others, simpler and more understandable.
EMI filter (10+)
High Frequency EMI Filter
The reason for the occurrence of high-frequency impulse noise is commonplace. The speed of light is not infinite, and the electromagnetic field propagates at the speed of light. When we have a device that somehow converts the mains voltage by frequent switching, we expect that in the power wires going to the network there will be ripple currents directed towards each other. On one wire, current flows into the device, on the other - it flows out. But it's not like that at all. Due to the finiteness of the field propagation velocity, the incoming current pulse is shifted in phase relative to the outgoing one. Thus, at a certain frequency, high-frequency currents in the network wires flow co-directionally, in phase.
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