Perhaps, after reading this article, you will not have to put the same size radiators on transistors.
Translation of this article.
A little note from the translator:
Firstly, in this translation there may be serious problems with the translation of terms, I did not study electrical and circuitry enough, but I still know something; I also tried to translate everything as clearly as possible, so I did not use such concepts as bootstrap, MOSFET, etc. Secondly, if spelling is already difficult to make a mistake (praise to word processors indicating errors), then making a mistake in punctuation is quite simple.
And on these two points, I ask you to kick me in the comments as hard as possible.
Now let's talk more about the topic of the article - with all the variety of articles on building various ground-based vehicles (cars) on MK, on Arduino, on<вставить название>, the circuit design itself, and even more so the motor connection diagram, is not described in sufficient detail. It usually looks like this:
- take the engine
- take components
- connect the components and the engine
- …
- PROFIT!1!
But building more complex circuits than simply spinning a PWM motor one way through an L239x usually requires knowledge of full bridges (or H-bridges), FETs (or MOSFETs), and drivers for them. If nothing limits, then p-channel and n-channel transistors can be used for a full bridge, but if the engine is powerful enough, then p-channel transistors will first have to be hung with a large number of radiators, then coolers should be added, but if it’s a pity to throw them away, then you can try other types of cooling, or just use only n-channel transistors in the circuit. But there is a small problem with n-channel transistors - it is sometimes quite difficult to open them “in a good way”.
So I was looking for something to help me with drawing the correct diagram, and I found an article on the blog of a young man called Syed Tahmid Mahbub. I decided to share this article.
In many situations, we must use FETs as high-level switches. Also, in many situations, we must use FETs as both upper and lower level switches. For example, in bridge circuits. In partial bridge circuits, we have 1 high level MOSFET and 1 low level MOSFET. In full bridge circuits we have 2 high level MOSFETs and 2 low level MOSFETs. In such situations, we will need to use both high and low level drivers together. The most common way to drive FETs in such cases is to use a low- and high-level switch driver for the MOSFET. Undoubtedly, the most popular driver IC is the IR2110. And in this article / tutorial I will talk about it.
You can download documentation for the IR2110 from the IR website. Here is the download link: http://www.irf.com/product-info/datasheets/data/ir2110.pdf
Let's first take a look at the block diagram, as well as the description and location of the pins:
Figure 1 - Functional block diagram of IR2110
Figure 2 - IR2110 Pinout
Figure 3 - Description of IR2110 pins
It's also worth mentioning that the IR2110 comes in two packages - a 14-pin PDIP for output mounting and a 16-pin SOIC for surface mounting.
Now let's talk about the various contacts.
VCC is low level power, must be between 10V and 20V. VDD is the logic power for the IR2110 and must be between +3V and +20V (with respect to VSS). The actual voltage you choose to use depends on the voltage level of the input signals. Here is the chart:
Figure 4 - Dependence of logical 1 on power
Usually a VDD of +5V is used. With VDD = +5V, the logic 1 input threshold is slightly higher than 3V. Thus, when VDD = +5V, IR2110 can be used to drive a load when input "1" is higher than 3 (something) volts. This means that the IR2110 can be used for almost all circuits, as most circuits are typically powered at around 5V. When you use microcontrollers, the output voltage will be higher than 4V (after all, the microcontroller often has VDD = +5V). When using an SG3525 or TL494 or other PWM controller, you will probably have to supply them with a voltage greater than 10V, which means that the outputs will be more than 8V at a logic one. Thus, the IR2110 can be used almost anywhere.
You can also lower VDD to around +4V if you are using a microcontroller or any chip that outputs 3.3V (eg dsPIC33). When designing circuits with the IR2110, I have noticed that sometimes the circuit does not work properly when the VDD of the IR2110 is set to less than +4V. Therefore, I do not recommend using VDD below +4V. In most of my circuits, the signal levels do not have voltages less than 4V as "1", and so I use VDD = +5V.
If for some reason in the circuit the logical "1" signal level has a voltage less than 3V, then you need to use a level converter / level translator, it will raise the voltage to acceptable limits. In such situations, I recommend stepping up to 4V or 5V and using the IR2110's VDD = +5V.
Now let's talk about VSS and COM. VSS is the land of logic. COM is the "low return" - basically the low level ground of the driver. It may look like they are independent, and one might think that it might be possible to isolate the driver outputs and the driver signal logic. However, this would be wrong. Although not internally connected, the IR2110 is a non-isolated driver, which means that VSS and COM must both be connected to ground.
HIN and LIN are logical inputs. A high signal on HIN means that we want to drive the top switch, that is, a high level output is performed on HO. A low signal on HIN means that we want to turn off the high-level MOSFET, that is, a low-level output is being made on HO. The output in HO, high or low, is not counted in relation to ground, but in relation to VS. We will soon see how the amplifying circuits (diode + capacitor) using VCC, VB and VS provide floating power to drive the MOSFET. VS is a floating power return. At a high level, the level on the HO is equal to the level on the VB, in relation to the VS. When low, the level at HO is equal to VS, relative to VS, effectively zero.
A high LIN signal means that we want to drive a low switch, that is, a high level output is performed on LO. A low LIN signal means that we want to turn off the low level MOSFET, that is, a low level output is performed on LO. The output in LO is considered relative to the ground. When the signal is high, the level in LO is the same as in VCC, relative to VSS, effectively ground. When the signal is low, the level in LO is the same as in VSS, relative to VSS, effectively zero.
SD is used as stop control. When the level is low, IR2110 is on - the stop function is disabled. When this pin is high, the outputs are disabled, disabling the control of the IR2110.
Now let's take a look at common configurations with the IR2110 to drive MOSFETs as high and low switches - half-bridge circuits.
Figure 5 - Basic circuit on IR2110 for half bridge control
D1, C1 and C2 together with IR2110 form an amplifying circuit. When LIN = 1 and Q2 is on, then C1 and C2 are charged to VB since one diode is below +VCC. When LIN = 0 and HIN = 1, the charge on C1 and C2 is used to add an extra voltage, VB in this case, above Q1's source level to drive Q1 in a high-switch configuration. A sufficiently large capacity must be chosen for C1 so that it is enough to provide the necessary charge for Q1 to keep Q1 on all this time. C1 should also not have too much capacitance, since the charging process will take a long time and the voltage level will not increase enough to keep the MOSFET on. The more time is required in the on state, the more capacity is required. Thus, a lower frequency requires a larger capacitance C1. A larger fill factor requires a larger capacitance C1. Of course, there are formulas for calculating capacitance, but for this you need to know many parameters, and we may not know some of them, for example, the leakage current of a capacitor. So I just estimated the approximate capacity. For low frequencies like 50Hz I use 47uF to 68uF capacitance. For high frequencies like 30-50kHz I use 4.7uF to 22uF capacitance. Since we are using an electrolytic capacitor, a ceramic capacitor must be used in parallel with this capacitor. A ceramic capacitor is optional if the amplifying capacitor is tantalum.
D2 and D3 discharge the gate of the MOSFETs quickly, bypassing the gate resistors and reducing turn-off time. R1 and R2 are current limiting gate resistors.
MOSV can be maximum 500V.
VCC must come from an uninterrupted source. You must install filter and decoupling capacitors from +VCC to ground for filtering.
Let's now look at some examples of circuits with IR2110.
Figure 6 - Diagram with IR2110 for a high voltage half bridge
Figure 7 - Diagram with IR2110 for a high-voltage full bridge with independent key management (clickable)
In Figure 7 we see an IR2110 used to drive a full bridge. There is nothing complicated in it, and I think you already understand this now. We can also apply a fairly popular simplification here: we connect HIN1 to LIN2, and we connect HIN2 to LIN1, thus we get control of all 4 keys using only 2 input signals, instead of 4, this is shown in Figure 8.
Figure 8 - Diagram with IR2110 for a high-voltage full bridge with key control with two inputs (clickable)
Figure 9 - Schematic with IR2110 as a high-voltage top-level driver
In Figure 9 we see the IR2110 used as the top level driver. The scheme is quite simple and has the same functionality as described above. One thing to consider - since we no longer have a low level switch, there must be a load connected from OUT to ground. Otherwise, the amplifying capacitor will not be able to charge.
Figure 10 - Schematic with IR2110 as a low-level driver
Figure 11 - Schematic with IR2110 as dual low level driver
If you're having trouble with the IR2110 and everything keeps crashing, burning or exploding, then I'm pretty sure it's because you're not using gate-source resistors, provided you've designed everything carefully, of course. NEVER FORGET ABOUT GATE-SOURCE RESISTORS. If you're interested, you can read about my experience with them here (I also explain the reason why resistors prevent damage).
Power MOSFETs and insulated gate bipolar transistors (IGBTs) are the basic elements of modern power electronics and are used as switching elements for high currents and voltages. However, to match low-voltage logic control signals with the gate drive levels of MOSFETs and IGBTs, intermediate matching devices are required - high-voltage drivers (hereinafter, for brevity, "high-voltage drivers" will mean "high-voltage drivers of MOSFETs and IGBTs").
In most cases, the following classification of high voltage drivers is used:
- Independent high-side and low-side half-bridge drivers integrated in a single chip ( High and Low Side Driver);
- High-side and low-side drivers connected in half-bridge ( Half Bridge Driver);
- High-side drivers ( high side driver);
- Low Shoulder Drivers ( low side driver).
On fig. 1 shows the control circuits corresponding to these types of drivers.
Rice. one.
In the first case (Fig. 1a), two independent loads are controlled from common control signals. Loads are respectively connected between the source of the lower transistor and the high-voltage supply rail (low-side driver), and between the drain of the upper transistor and ground (high-side driver). The so-called midpoints (the drain of the upper transistor and the source of the lower transistor) are not connected to each other.
In the second case (Fig. 1b), the middle points are connected. Moreover, the load can be switched on both on the upper and on the lower arm, but connected to the midpoint in the same way as a half-bridge circuit (the so-called full bridge circuit). Strictly speaking, in scheme 1a, nothing prevents you from connecting the midpoints. But in this case, with a certain combination of input signals, it is possible to simultaneously open two transistors at once and, accordingly, an excessively large current flows from the high-voltage bus to the ground, which will lead to the failure of one or both transistors at once. The exception of such a situation in this scheme is the concern of the developer. In half-bridge drivers (scheme 1b), such a situation is excluded at the level of the internal control logic of the microcircuit.
In the third case (1c), the load is connected between the drain of the upper transistor and ground, and in the fourth case (1d), between the source of the lower transistor and the high-voltage power bus, i.e. two "halves" of scheme 1a are implemented separately.
In recent years, STMicroelectronics has focused (in the niche of high-voltage drivers) only on drivers of the first two types (families L638x And l639x, which will be discussed below). However, earlier designs contain driver ICs that control whether a single MOSFET or IGBT transistor turns on or off (the “Single” category in STMicroelectronics terms). With a certain switching scheme, these drivers can control the load of both the upper and lower shoulders. We also note the microcircuit TD310 - three independent single drivers in one package. Such a solution will be effective in managing a three-phase load. STMicroelectronics classifies this chip as a driver in the "Multiple" category.
L368x
Table 1 lists the composition and parameters of the L368x family chips. The microcircuits of this family include both independent high- and low-side (H&L) drivers and half-bridge (HB) drivers.
Table 1. L638x Family Driver Parameters
| Name | Voffset, V | Io+, mA | Io-, mA | Ton, ns | Toff, ns | Tdt, ns | Type | Control |
|---|---|---|---|---|---|---|---|---|
| L6384E | 600 | 400 | 650 | 200 | 250 | Prog. | HB | IN/SD |
| L6385E | 600 | 400 | 650 | 110 | 105 | H&L | HIN/LIN | |
| L6386E | 600 | 400 | 650 | 110 | 150 | H&L | HIN/LIN/SD | |
| L6387E | 600 | 400 | 650 | 110 | 105 | H&L | HIN/LIN | |
| L6388E | 600 | 200 | 350 | 750 | 250 | 320 | HB | HIN/LIN |
Let's explain some parameters:
V OFFSET - the maximum possible voltage between the source of the upper transistor and ground;
I O+ (I O-) - maximum output current with the upper (lower) transistor of the output stage of the microcircuit open;
T ON (T OFF) - signal propagation delay from the HIN and LIN inputs to the HO and LO outputs when turned on (off);
T DT - pause time - a parameter related to half-bridge drivers. When changing active states, the logic circuit forcibly introduces pauses to avoid turning on the upper and lower arms at the same time. For example, if the lower arm turns off, then both arms are turned off for some time and only then the upper one turns on. And vice versa, if the upper arm is switched off, then both arms are switched off for some time and then the lower one is switched on. This time can either be fixed (as in L6388E), or be set by selecting the value of the corresponding external resistor (as in L6384E).
Control. Chips of independent drivers of the upper and lower shoulders are controlled by the HIN and LIN inputs. Moreover, the high level of the logical signal turns on, respectively, the upper or lower shoulder of the driver. In addition, the L6386E chip uses an additional SD input, which disables both arms, regardless of the state at the HIN and LIN inputs.
The L6384E chip uses SD and IN signals. The SD signal disables both arms, regardless of the state at the IN input. Signal IN = 1 is equivalent to a combination of signals (HIN = 1, LIN = 0) and vice versa, IN = 0 is equivalent to a combination of signals (HIN = 0, LIN = 1). Thus, the simultaneous switching on of the transistors of the upper and lower arms is impossible in principle.
In the L6388E chip, control is carried out by the HIN and LIN inputs, so it is fundamentally possible to apply a combination (HIN = 1, LIN = 1) to the inputs, however, the internal logic circuit converts it into a combination (HIN = 0, LIN = 0), thus eliminating , simultaneously turning on both transistors.
As for the parameters, let's start with H&L type chips.
The value of V OFFSET, equal to 600 Volts, is in a sense the standard for microcircuits of this class.
The output current I O+ (I O-) of 400/650 mA is an average value for typical general purpose transistors. Compared to the IRS family (G5 HVIC generation), International Rectifier mainly offers 290/600mA ICs. However, the International Rectifier line also includes models with 2500/2500 mA parameters (IRS2113) and slightly slower performance or microcircuits with output currents up to 4000/4000 mA (IRS2186). However, in this case, the switching time compared to L6385E increases to a value of 170/170 ns.
Switching time. The T ON (T OFF) values of 110/105 ns (for the L6385E) exceed those of the IRS family of microcircuits (albeit not very significantly). The best performance (60/60 ns) was achieved by International Rectifier in the IRS2011 model, but at the expense of reducing the VOFFSET voltage to 200 V.
However, we note that STMicroelectronics offers drivers in which the common wire of the input (low-voltage) and output (high-voltage) stages is a single one. International Rectifier, in addition to chips with a similar architecture, offers drivers with separate common buses for the input and output stages.
Comparing the parameters of the L6384E half-bridge driver with International Rectifier products, we can conclude that it is inferior (both in output currents and in speed) only to the IRS21834 model, which implements the HIN / -LIN input logic. If IN/-SD input logic is critical, then the L6384E outperforms International Rectifier products.
Let's take a closer look at the L6385E driver chip, the structure and switching circuit of which is shown in fig. 2.
Rice. 2.
The microcircuit contains two independent drivers of the upper (HVG output) and lower side (LVG output). The implementation of the low-side driver is fairly trivial, since the potential at the GND pin is constant and, therefore, the task is to convert the input low-voltage logic signal LIN to the voltage level at the LVG output necessary to turn on the low-side transistor. In the upper side, the potential at the OUT pin changes depending on the state of the lower transistor. There are various circuit solutions used to build the upper-side cascade. In this case, a relatively simple and inexpensive bootstrap control circuit (a circuit with a "floating" power supply) is used. In such a scheme, the duration of the control pulse is limited by the value of the bootstrap capacitance. In addition, it is necessary to provide conditions for its constant charge using a high-voltage high-speed level shift cascade. This stage provides the conversion of logic signals to the levels necessary for stable operation of the high-side transistor control circuit.
When the control voltage drops below a certain limit, the output transistors can go into a linear mode of operation, which, in turn, will lead to overheating of the crystal. To prevent this, voltage monitoring circuits (UVLO - Under Voltage Lock Out) and for the upper (potential control V BOOT) and lower (potential control V CC) arms.
Modern high voltage drivers tend to integrate the bootstrap diode into the integrated circuit package. This eliminates the need for an external diode, which is quite bulky compared to the driver chip itself. The built-in bootstrap diode (more precisely, the bootstrap circuit) is used not only in the L6385E driver, but also in all other microcircuits of this family.
The L6386E is a variant of the L6385E with additional features. Its structure and switching circuit are shown in fig. 3.
Rice. 3.
Main differences between L6386E and L6385E. First, an additional SD input has been added, a low signal level on which turns off both transistors, regardless of the state of the HIN and LIN inputs. Often used as an emergency shutdown signal that is not associated with the input control signal generation circuit. Secondly, a cascade has been added to control the current flowing through the transistor of the lower cascade. Comparing with the previous circuit, we see that the drain of the low-side transistor is connected to the ground not directly, but through a current resistor (current sensor). If the voltage drop across it exceeds the threshold value V REF , then a low level is generated at the DIAG output. Note that this state does not affect the operation of the circuit, but is only an indicator.
A few words about the use of chips of the L638x family. The limited space of this article does not allow for application examples, but STMicroelectronics' L638xE Application Guide provides examples of a three-phase motor control circuit, a dimmable fluorescent lamp ballast circuit, DC/DC converters with various architectures, and a number of others. There are also diagrams of demo boards for all microcircuits of this family (including the topology of printed circuit boards).
Summing up the analysis of the L638x family, we note that without having unique characteristics in terms of some individual parameters, the drivers of this family are among the best in the industry both in terms of the combination of parameters and the applied technical solutions.
High voltage driver family
half-bridge circuit L639x
At first glance, microcircuits of this family can be considered a development of the L6384E microcircuit. However, analyzing the functionality of the L639x family of drivers, it is very difficult to recognize the L6384E as a prototype (except perhaps due to the lack of other half-bridge drivers in the STMicroelectronics line). Table 2 lists the composition and parameters of the L639x family chips.
Table 2. L639x Family Driver Parameters
| Name | Voffset, V | Io+, mA | Io-, mA | Ton, ns | Toff, ns | Tdt, ms | Type | Smart SD | OU | Comp. | Control |
|---|---|---|---|---|---|---|---|---|---|---|---|
| L6390 | 600 | 270 | 430 | 125 | 125 | 0,15…2,7 | HB | eat | eat | eat | HIN/-LIN/-SD |
| L6392 | 600 | 270 | 430 | 125 | 125 | 0,15…2,7 | HB | eat | HIN/-LIN/-SD | ||
| L3693 | 600 | 270 | 430 | 125 | 125 | 0,15…2,7 | HB | eat | PH/-BR/-SD |
The main feature of microcircuits of this family is the presence of additional built-in elements: an operational amplifier or comparator (for L6390- both of those). On fig. 4 shows the structure and switching circuit of the L6390 chip.
Rice. 4.
What advantages do additional elements provide in practical applications? Operational amplifiers (in the L6390 and L6392) are designed to measure the current flowing through the load. Moreover, since both outputs are available (OP + and OP-), it becomes possible to form both an absolute value and a deviation from a certain reference voltage (corresponding, for example, to the maximum allowable value) at the corresponding output of the microcircuit. In the L6390 driver, the comparator performs a very specific "intelligent shutdown" function ( smart shutdown) — i.e. when the maximum allowable current in the load is exceeded, the comparator begins to influence the logic of the driver and provides a smooth shutdown of the load. The shutdown speed is set by the RC circuit connected to the SD/OD pin. Moreover, since this output is bidirectional, it can be both an error indication output for the control microcontroller and an input for forced shutdown.
All microcircuits contain protection logic against simultaneous opening of the transistors of the upper and lower shoulders and, accordingly, the formation of a pause when the output state changes. The pause time T DT for all microcircuits of the family is programmable and is determined by the value of the resistor connected to the DT pin.
Control logic in L6390 and L6392 the same type - HIN, LIN and SD signals.
Chip difference L6393 from the L6390 and L6392 is not only the lack of an operational amplifier. The comparator in the L6393 is independent of the rest of the circuit and, in principle, can be used for arbitrary purposes. However, the most reasonable application is current control and the formation of an excess sign (similar to the DIAG pin in the L6386E chip discussed above). The main difference lies in the control logic - the combination of PHASE, BRAKE and SD control signals is quite rare (if not unique) for microcircuits of this class. The control sequence diagram is shown in fig. five.
Rice. five.
The cyclogram is focused on control directly from motor signals, for example, direct current and implements the so-called. delayed stop mechanism. Let's assume that BRAKE is a signal to the actuator, i.e. its low level turns on the motor regardless of the state of the PHASE signal. Again, assume that PHASE is a signal from a feedback sensor, such as a frequency sensor mounted on the motor shaft, or a limit sensor indicating a break point. Then the high level of the BRAKE signal will stop the engine immediately, but only on the positive edge of the PHASE signal. For example, if we are talking about the drive of the carriage, then the stop signal (high level BRAKE) can be given in advance, but the stop will occur only at a specific point (when the PHASE sensor is triggered).
On fig. 6 shows the structure and switching circuit of the L6393 chip.
Rice. 6.
About parameters. The values of the output currents I O+ (I O-), equal to 270/430 mA, are inferior to the International Rectifier microcircuits (which, as noted above, are typical 290/600 mA). Nevertheless, the dynamic parameters T ON /T OFF (125/125 ns) are superior (and often significantly) to all IRS family chips.
Conclusions on the L639x family. With sufficiently high quantitative characteristics, which in itself allows us to classify the L639x family as an industry leader, additional functions give a qualitative leap, since they allow implementing in one microcircuit those functions that were previously implemented using a number of additional components.
Conclusion
Of course, the range of high-voltage drivers from STMicroelectronics cannot be considered very wide (at least in comparison with similar products from International Rectifier). Nevertheless, the quantitative and qualitative characteristics of the considered families are not inferior to the best IR products.
Speaking of MOSFET and IGBT drivers, one cannot but mention the transistors themselves; STMicroelectronics produces a fairly wide range of field-effect (for example, MDMESH V and SuperMesh3) and insulated gate bipolar transistors. Since these electronic components were recently covered in this journal, they are left outside the scope of this article.
Finally, as mentioned above, STMicroelectronics' line of MOSFET and IGBT drivers is not limited to half-bridge drivers. The nomenclature of drivers of the "Single" and "Multiple" categories and their parameters can be found on the official website of STMicroelectronics - http://www.st.com/ .
Literature
1. L638xE Application Guide// ST Microelectronics document an5641.pdf.
2. Yachmennikov V. Increasing efficiency with MDmesh V transistors // News of Electronics, No. 14, 2009.
3. Ilyin P., Alimov N. Review of STMicroelectronics MOSFET and IGBT// Electronics News, No. 2, 2009.
4. Medzhahed D. Highly efficient solutions based on SuperMESH3 transistors // News of Electronics, No. 16, 2009.
MDMEDH V in PowerFlat package
STMicroelectronics, a world leader in power MOSFETs, has developed a new, high-performance PowerFlat package for the MDMESH V family of transistors, specifically designed for surface mounting. Case dimensions 8x8 mm with a height of 1 mm (PowerFlat 8x8 HV). Its low height allows you to create thinner power supplies, as well as reduce the size of the printed circuit board or increase the density of the installation. The drain contact in the PowerFlat package is a large, exposed metal surface, which improves heat dissipation and thus improves reliability. This housing is capable of operating in the temperature range of -55…150°C.
Transistors of the MDMESH V family are the best transistors in the world in terms of open channel resistance in the operating voltage range of 500 ... 650 V. For example, transistors of the series STW77N65M5 from the MDMESH V family have a maximum Rdson value of 0.033 Ohm for an operating voltage of 650 V and a maximum static current of 69 A. At the same time, the gate charge of such a transistor is only 200 nK. STL21N65M5 - it is the first transistor from the MDMESH V family in a PowerFlat package. With an operating voltage of 650 V, the STL21N65M5 transistor has an open channel resistance of 0.190 ohms and a maximum static current of 17 A, while its gate charge is 50 nK.
About ST Microelectronics
The driver is a power amplifier and is intended for direct control of the power switch (sometimes keys) of the converter. It must amplify the control signal in terms of power and voltage and, if necessary, provide its potential shift.
The output node of the insulated gate driver (MOSFET, IGBT) must meet the following requirements:
MIS transistors and IGBTs are voltage controlled devices, however, in order to increase the input voltage to the optimum level (12-15 V), it is necessary to provide an appropriate charge in the gate circuit.
To limit the rate of current rise and reduce dynamic noise, it is necessary to use series resistances in the gate circuit.
Drivers for controlling complex converter circuits contain a large number of elements, so they are produced in the form of integrated circuits. These microcircuits, in addition to power amplifiers, also contain level conversion circuits, auxiliary logic, delay circuits for forming “dead” time, as well as a number of protections, for example, from overcurrent and short circuit, supply voltage reduction and a number of others. Many companies produce numerous functional range: low-key bridge drivers, high-key bridge drivers, high- and low-key drivers with independent control of each of them, half-bridge drivers, which often have only one control input and can be used for a symmetrical control law, drivers to drive all transistors in the bridge circuit.
A typical circuit for switching on the driver of the upper and lower switches from International Rectifier IR2110 with a bootstrap power supply principle is shown in Fig. 3.1, a. The control of both keys is independent. The difference between this driver and others is that the IR2110 has an additional level conversion circuit both in the lower and upper channels, which allows you to separate the power supply of the microcircuit logic from the driver supply voltage by level. It also contains protection against undervoltage supply to the driver and a high-voltage "floating" source.
Capacitors C D, C C are designed to suppress high-frequency interference in the logic and driver power circuits, respectively. A high-voltage floating source is formed by a capacitor C1 and a diode VD1 (bootstrap power supply).
The driver outputs are connected to power transistors using gate resistors R G1 and R G2.
Since the driver is built on field elements and the total power consumed for control is insignificant, capacitor C1 can be used as a power source for the output stage, recharged from the power supply U PIT through a high-frequency diode VD1. Capacitor C1 and diode VD1 together form a high-voltage "floating" power source designed to control the upper transistor VT1 of the bridge rack. When the lower transistor VT2 conducts current, the source of the upper transistor VT1 is connected to a common power wire, the diode VD1 opens and the capacitor C1 is charged to a voltage U C1 \u003d U PIT - U VD1. On the contrary, when the lower transistor goes into the closed state and the upper transistor VT2 starts to open, the diode VD1 is backed up by the reverse voltage of the power supply. As a result, the output stage of the driver begins to be powered exclusively by the discharge current of the capacitor C1. Thus, the capacitor C1 constantly "walks" between the common wire of the circuit and the wire of the power supply (point 1).
When using the bootstrap-powered IR2110 driver, special attention should be paid to the selection of high-voltage "floating" source elements. The VD1 diode must withstand a large reverse voltage (depending on the power supply of the circuit), the allowable forward current is approximately 1 A, the recovery time t rr \u003d 10-100 ns, i.e. be fast. The literature recommends the SF28 diode (600 V, 2 A, 35 ns), as well as the diodes UF 4004…UF 4007, UF 5404…UF 5408, HER 105… HER 108, HER 205…HER 208 and other “ultra-fast” classes .
The driver circuit is designed in such a way that a high logical signal level at any HIN and LIN input corresponds to the same level at its HO and LO output (see Fig. 3.1 b, common-mode driver). The appearance of a high level of a logic signal at the SD input leads to the blocking of the transistors of the bridge rack.
It is advisable to use this microcircuit to control the inverter keys with PWM output voltage regulation. At the same time, it must be remembered that time delays (“dead” time) must be provided for in the control system in order to prevent through currents when switching transistors of the bridge rack (VT1, VT2 and VT3, VT4, Fig. 1.1).
Capacitance C1 is a bootstrap capacitance, the minimum value of which can be calculated by the formula:
where Q 3 - the value of the gate charge of a powerful key (reference value);
I Pete- current consumption of the driver in static mode (reference value, usually I Pete ≈ I G c T powerful key)
Q 1 - cyclic change in the charge of the driver (for 500-600 - volt drivers 5 nK);
V P– supply voltage of the driver circuit;
– voltage drop across the bootstrap diode VD1;
T– switching period of powerful keys.
Fig.3.1. Typical circuit for switching on the IR2110 driver (a) and timing diagrams of its signals at the inputs and outputs (b)
V DD - power supply for the logic of the microcircuit;
V SS - common point of the logical part of the driver;
HIN, LIN - logical input signals that control the upper and lower transistors, respectively;
SD – logic input of driver disable;
V CC - driver supply voltage;
COM is the negative pole of the power supply V CC ;
HO, LO - driver output signals that control the upper and lower transistors, respectively;
V B is the supply voltage of the high-voltage "floating" source;
V S is the common point of the negative pole of the high-voltage "floating" source.
The obtained value of the bootstrap capacitance must be increased by a factor of 10-15 (usually C in the range of 0.1-1 μF). This should be a high-frequency capacitance with a low leakage current (ideally, tantalum).
Resistors R G 1, R G 2 determine the turn-on time of powerful transistors, and diodes VD G 1 and VD G 2, by shunting these resistors, reduce the turn-off time to minimum values. Resistors R 1, R 2 have a small value (up to 0.5 Ohm) and equalize the spread of ohmic resistances along the common control bus (mandatory if a powerful key is a parallel connection of less powerful transistors).
When choosing a driver for power transistors, consider:
Power transistor control law:
For symmetrical law, high and low key drivers and half-bridge drivers are suitable;
Asymmetric law requires high and low key drivers with independent control of each powerful key. For asymmetric law, drivers with transformer galvanic isolation are not suitable.
Powerful key parameters (I to or I drain).
The approximate approach is usually used:
I out dr max =2 A can control a powerful VT with a current of up to 50 A;
I out dr max \u003d 3 A - control a powerful VT with a current of up to 150 A (otherwise, the turn-on and turn-off time increases significantly and the power losses for switching increase), i.e. a high-quality transistor with an erroneous choice of driver loses its main advantages.
Accounting for additional functions.
Firms produce drivers with numerous service functions:
Various powerful key protections;
Driver undervoltage protection;
With built-in bootstrap diodes;
With adjustable and non-adjustable delay time for turning on a powerful VT in relation to the moment of turning off another one (combating through currents in a half-bridge);
With built-in or without galvanic isolation. In the latter case, at the input of the driver, it is necessary to connect a galvanic isolation microcircuit (most often, a high-frequency diode optocoupler);
In-phase or anti-phase;
Power supply for drivers (bootstrap type of power supply or three galvanically isolated power supplies are required).
With the equivalence of several types of drivers, preference should be given to those that switch the gate current of powerful transistors using bipolar VTs. If this function is performed by field-effect transistors, then there may be failures in the driver under certain circumstances (overloads) due to the “latching” trigger effect.
After choosing the type of driver (and its data), measures are needed to combat through currents in the half-bridge. The standard way is to turn off a powerful key instantly, and turn on a locked one with a delay. For this purpose, diodes VD G 1 and VD G 2 are used, which, when closing VT, shunt the gate resistors, and the shutdown process will be faster than unlocking.
In addition to shunting the gate resistors R G 1 and R G 2 using diodes (VD G 1, VD G 2, Fig. 3.1), to combat through currents in the P-circuit of a powerful cascade, firms produce integrated drivers that are asymmetric in terms of the output turn-on current VT I dr out m ah on and off I dr out m ah off(for example I dr out m ah on=2A, I dr out m ah off=3A). This sets the asymmetric output resistances of the microcircuit, which are connected in series with the gate resistors R G 1 and R G 2 .
,
.
where all the values in the formulas are the reference data of a particular driver.
For a symmetrical (by currents) driver, the equality
.
So, to prevent the occurrence of through currents, it is necessary to select the total resistance value in the gate circuit (due to 
,
and, accordingly, by adjusting the charge current of the gate capacitance VT), turn-on delay 
transistor greater than or equal to the time taken to close VT
where 
– drain current decay time (reference value);
is the delay time of the start of switching off VT with respect to the moment when the blocking voltage is applied to the gate, which depends on the magnitude of the gate discharge current (respectively, it depends on the total resistance in the gate circuit). With shunt gate diodes (VD G 1, VD G 2, Fig. 3.1), the discharge current is uniquely determined by the resistance 
. Therefore, to determine 
solve the following proportion
(corresponding) - 
(corresponding) - 
If the corrected value 
will be much more 
, then this indicates an incorrect choice of the type of driver in terms of power (large 
) and this corrects for the worse the performance of powerful keys. For the final determination of the value 
you can use the technical reference data of the powerful VT. For this, the proportion
(corresponding) - 
(corresponding) - 
(If the solution gives a negative value R G 1, then the turn-on delay will be provided with a margin by the output impedance of the driver).
To facilitate the fight against through currents, some manufacturers already at the manufacturing stage ensure that t off< t вкл (например, сборка – полумост СМ35084-5F фирмы Mitsubishi Elektric с динамическими параметрами: t з вкл =1,1 мс, t вкл =2,4 мс, t з выкл =0,9 мс, t выкл =0,5 мс).
Diodes VD G 1 and VD G 2 must be high-frequency and withstand the supply voltage of the driver with a margin.
To combat through currents (for a symmetrical control law), you can select the desired half-bridge driver (if it is suitable for other parameters), whose delay time is adjustable in the range of 0.4 ... 5 μs (for example, IR drivers such as IR2184 or IR21844), if their delay is greater than or equal to t off.
In conclusion, it is worth noting that instead of old driver modifications, firms release new types that are compatible with the old ones, but may have additional service functions (usually built-in bootstrap diodes, or rather, bootstrap transistors that perform the function of diodes that were previously absent). For example, the IR2011 driver has been discontinued and a new IRS2011 or IR2011S has been introduced to replace it (an ambiguous entry in various manuals).
The article is devoted to the developments of LLC "Electrum AV" for industrial use, according to their characteristics, similar to modular devices manufactured by Semikron and CT Concept.
Modern concepts for the development of power electronics, the level of the technological basis of modern microelectronics determine the active development of systems built on IGBT devices of various configurations and powers. In the state program "National Technological Base", two works are devoted to this area on the development of a series of medium power IGBT modules at the Kontur enterprise (Cheboksary) and a series of high power IGBT modules at the Kremniy enterprise (Bryansk). At the same time, the use and development of systems based on IGBT modules is limited by the lack of domestic driver devices for controlling IGBT gates. This problem is also relevant for high-power field-effect transistors used in converter systems with voltages up to 200 V.
Currently, on the Russian "electronic" market, control devices for high-power field-effect and IGBT transistors are represented by Agilent Technologies, IR, Powerex, Semikron, CT Concept. IR and Agilent products contain only a transistor control signal conditioner and protective circuits and require additional elements when working with transistors of high power or at high frequencies for their application: a DC / DC converter of the necessary power to form the supply voltages of output stages, powerful external output stages for generating gate control signals with the required edge steepness, protective elements (zener diodes, diodes, etc.), interface elements of the control system (input logic, formation of a control diagram for half-bridge devices, optically isolated status signals of the state of the controlled transistor, supply voltages etc.). Powerex products also require a DC/DC converter, while TTL, CMOS, and fiber optics require additional external components. Also, there are no necessary status signals with galvanic isolation.
The most functionally complete are Semikron's (SKHI series) and CT Concept's (Standard or SCALE types). CT Concept drivers of the Standart series and SKHI drivers are made in the form of printed circuit boards with connectors for connecting to the control system and controlled transistors with the necessary elements installed on them and with the possibility of installing tuning elements by the consumer. In terms of their functional and parametric features, the products are close.
The nomenclature of SKHI drivers is shown in Table 1.
Table 1. Nomenclature of SKHI drivers
| Semikron driver type | Number of channels | Max voltage on control. transistor tore, V | Gate voltage change, V | Max imp. out. current, A | Max gate charge, µC | Frequency, kHz | Insulation voltage, kV | DU/dt, kV/µs |
| SKHI 10/12 | 1 | 1200 | +15/–8 | 8 | 9,6 | 100 | 2,5 | 75 |
| SKHI 10/17 | 1 | 1700 | +15/–8 | 8 | 9,6 | 100 | 4 | 75 |
| SKHI 21A | 1 | 1200 | +15/–0 | 8 | 4 | 50 | 2,5 | 50 |
| SKHI 22A/22V | 2 | 1200 | +15/–7 | 8 | 4 | 50 | 2,5 | 50 |
| SKHI 22A/H4 | 2 | 1700 | +15/–7 | 8 | 4 | 50 | 4 | 50 |
| SKHI 22V/H4 | 2 | 1700 | +15/–7 | 8 | 4 | 50 | 4 | 50 |
| SKHI 23/12 | 2 | 1200 | +15/–8 | 8 | 4,8 | 100 | 2,5 | 75 |
| SKHI 23/17 | 2 | 1700 | +15/–8 | 8 | 4,8 | 100 | 4 | 75 |
| SKHI 24 | 2 | 1700 | +15/–8 | 8 | 5 | 50 | 4 | 50 |
| SKHI 26W | 2 | 1600 | +15/–8 | 8 | 10 | 100 | 4 | 75 |
| SKHI 26F | 2 | 1600 | +15/–8 | 8 | 10 | 100 | 4 | 75 |
| SKHI 27W | 2 | 1700 | +15/–8 | 30 | 30 | 10 | 4 | 75 |
| SKHI 27F | 2 | 1700 | +15/–8 | 30 | 30 | 10 | 4 | 75 |
| SKHI 61 | 6 | 900 | +15/–6,5 | 2 | 1 | 50 | 2,5 | 15 |
| SKHI 71 | 7 | 900 | +15/–6,5 | 2 | 1 | 50 | 2,5 | 15 |
| SKHIBS 01 | 7 | 1200 | +15/–8 | 1,5 | 0,75 | 20 | 2,5 | 15 |
CT Concept's SCALE drivers are based on a basic hybrid assembly and include the main elements for driving high-power field-effect or IGBT transistors, which are mounted on a printed circuit board, with the ability to install the necessary tuning elements. The board is also equipped with the necessary connectors and sockets.
The range of basic hybrid SCALE driver assemblies from CT Concept is shown in Table 2.
Driver devices manufactured by Electrum AV are completely complete, functionally complete devices containing all the necessary elements for controlling the gates of powerful transistors, providing the necessary levels of matching of current and potential signals, rise times and delays, as well as the necessary levels of protection for controlled transistors at dangerous levels of saturation voltage (current overload or short circuit) and insufficient gate voltage. The used DC/DC converters and transistor output stages have the necessary power to ensure switching of controlled transistors of any power with sufficient speed to ensure minimal switching losses. DC/DC converters and optocouplers have sufficient levels of galvanic isolation for high voltage applications.
Table 2. Nomenclature of basic hybrid assemblies of SCALE drivers from CT Concept
| CT Concept driver type | Number of channels | Supply voltage driver-faith, V | Max imp. output current, A | Max voltage on ex. transistor tore, V | Output power, W | Delay, ns | Isolation voltage, V | du/dt, kV/µs | entrance |
| IGD 508E | 1 | ±15 | ±8 | 3300 | 5 | 225 | 5000 | Vols | |
| IGD 515E | 1 | ±15 | ±15 | 3300 | 5 | 225 | 5000 | Vols | |
| IGD 608E | 1 | ±15 | ±8 | 1200 | 6 | 60 | 4000 | >50 | Trance |
| IGD608A1 17 | 1 | ±15 | ±8 | 1700 | 6 | 60 | 4000 | >50 | Trance |
| IGD 615A | 1 | ±15 | ±15 | 1200 | 6 | 60 | 4000 | >50 | Trance |
| IGD615A1 17 | 1 | ±15 | ±15 | 1700 | 6 | 60 | 4000 | >50 | Trance |
| IHD 215A | 2 | ±15 | ±1.5 | 1200 | 1 | 60 | 4000 | >50 | Trance |
| IHD 280A | 2 | ±15 | ±8 | 1200 | 1 | 60 | 4000 | >50 | Trance |
| IHD280A1 17 | 2 | ±15 | ±8 | 1700 | 1 | 60 | 4000 | >50 | Trance |
| IHD 680A | 2 | ±15 | ±8 | 1200 | 3 | 60 | 4000 | >50 | Trance |
| IHD680A1 17 | 2 | ±15 | ±8 | 1700 | 3 | 60 | 4000 | >50 | Trance |
| IHD580F | 2 | ±15 | ±8 | 2500 | 2,5 | 200 | 5000 | Vols |
This article will present devices MD115, MD150, MD180 (MD115P, MD150P, MD180P) for controlling single transistors, as well as MD215, MD250, MD280 (MD215P, MD250P, MD280P) for controlling half-bridge devices.
Single-channel IGBT driver module and powerful field-effect transistors: MD115, MD150, MD180, MD115P, MD150P, ID180P
The driver module MD115, MD150, MD180, MD115P, MD150P, MD180P is a hybrid integrated circuit for controlling IGBTs and high-power field-effect transistors, including in the case of their parallel connection. The module provides matching of current and voltage levels with most IGBTs and powerful field-effect transistors with a maximum allowable voltage of up to 1700 V, protection against overload or short circuit, against insufficient voltage at the transistor gate. The driver generates an "accident" signal in case of a violation of the transistor's operating mode. With the help of external elements, the driver's operating mode is configured for optimal control of different types of transistors. The driver can be used to drive transistors with "Kelvin" outputs or to control current with a current sense resistor. Devices MD115P, MD150P, MD180P contain a built-in DC / DC converter to power the output stages of the driver. Devices MD115, MD150, MD180 require an external isolated power supply.
Pin assignment
1 - "emergency +" 2 - "emergency -" 3 - "input +" 4 - "input -" 5 - "U supply +" (only for models with index "P") 6 - "U supply -" (only for models with the index “P”) 7 - “Common” 8 - “+E supply” 9 - “output” - transistor gate control 10 - “–E supply” 11 - “for example” - input for monitoring the saturation voltage of the controlled transistor 12 - "current" - input to control the current flowing through the controlled transistor
IA215, IA250, IA280, IA215I, IA250I, IA280I dual-channel IGBT and power FET driver modules
Driver modules MD215, MD250, MD280, MD215P, MD250P, MD280P - a hybrid integrated circuit for controlling IGBTs and powerful field-effect transistors in two channels, both independently and in half-bridge connection, including when transistors are connected in parallel. The driver provides matching of current and voltage levels with most IGBTs and powerful field-effect transistors with maximum allowable voltages up to 1700 V, protection against overloads or short circuits, insufficient voltage at the transistor gate. The driver inputs are galvanically isolated from the power section with an isolation voltage of 4 kV. The driver contains internal DC/DC converters that form the necessary levels to control the gates of transistors. The device generates the necessary status signals characterizing the mode of operation of the transistors, as well as the presence of power. With the help of external elements, the driver's operating mode is configured for optimal control of different types of transistors.
Table 4. Designation of the pins of the dual-channel IGBT driver module and power field-effect transistors
| Pin No. | Designation | Function | Pin No. | Designation | Function |
| 14 | ВХ1 "+" | Direct control input of the first channel | 15 | IR | Measuring collector for monitoring the saturation voltage on the controlled transistor of the first channel |
| 13 | BX1 "-" | Inverted control input of the first channel | 16 | SG1 | Saturation voltage control input with adjustable threshold and blocking time of the first channel |
| 12 | ST "+ E pit" | Status of the supply voltage of the output stage of the first channel | 17 | Out2 | Transistor gate control output with adjustable turn-on time of the controlled transistor of the first channel |
| 11 | Sz | Input for connecting an additional capacitor (setting the turn-on delay time) of the first channel | 18 | Out1 | Transistor gate control output with adjustable turn-off time of the controlled transistor of the first channel |
| 10 | ST | Alarm status output on the controlled transistor of the first channel | 19 | –E pit | |
| 9 | BLOCK | Interlock input | 20 | Common | Supply voltage outputs of the power part of the driver of the first channel |
| 8 | Not involved | 21 | +E pit | Supply voltage outputs of the power part of the driver of the first channel | |
| 7 | +5V | 22 | + E pit " | ||
| 6 | Input for connecting the power supply of the input circuit | 23 | Common" | Supply voltage outputs of the power part of the driver of the second channel | |
| 5 | BX2 "+" | Direct control input of the second channel | 24 | –E pit " | Supply voltage outputs of the power part of the driver of the second channel |
| 4 | BX2 "-" | Inverted control input of the second channel | 25 | Out1" | Transistor gate control output with adjustable turn-on time of the controlled transistor of the second channel |
| 3 | ST "+ E pit" 9 | Status of the supply voltage of the output stage of the second channel | 26 | Out2" | Transistor gate control output with adjustable turn-off time of the controlled transistor of the second channel |
| 2 | Sz9 | Input for connecting an additional capacitor (setting the switching delay time) of the second channel | 27 | IK1" | Saturation voltage control input with adjustable threshold and blocking time of the second channel |
| 1 | ST9 | Alarm status output on the controlled transistor of the second channel | 28 | IR" | Measuring collector for monitoring the saturation voltage on the controlled transistor of the second channel |
Devices of both types МД1ХХХ and МД2ХХХ ensure the formation of control signals for transistor gates with separately adjustable charging and discharging currents, with the required dynamic parameters, provide voltage control and protection of transistor gates in case of insufficient or excessive voltage on them. Both types of devices monitor the saturation voltage of the controlled transistor and perform a smooth emergency shutdown of the load in critical situations, generating an optocoupler isolation signal indicating this. In addition to these functions, devices of the MD1XXX series have the ability to control the current through a controlled transistor using an external current-measuring resistor - a "shunt". Such resistors, with resistances from 0.1 to several mOhm and powers of tens and hundreds of watts, made on ceramic bases in the form of nichrome or manganin strips of precise geometry with adjustable ratings, were also developed by Electrum AV LLC. More detailed information about them can be found on the website www.orel.ru/voloshin.
Table 5. Main electrical parameters
| input circuit | ||||
| min. | type. | Max. | ||
| Supply voltage, V | 4,5 | 5 | 18 | |
| Consumption current, mA | no more than 80 without load no more than 300mA with load | |||
| Input logic | CMOS 3 -15 V, TTL | |||
| Control input current, mA | no more than 0.5 | |||
| Output voltage st, V | no more than 15 | |||
| Output current on output st, mA | at least 10 | |||
| output circuit | ||||
| Peak output current, A | ||||
| MD215 | no more than 1.5 | |||
| MD250 | no more than 5.0 | |||
| MD280 | no more than 8.0 | |||
| Output average current, mA | no more than 40 | |||
| Maximum switching frequency, kHz | at least 100 | |||
| Rate of voltage change, kV/μs | at least 50 | |||
| Maximum voltage on the controlled transistor, V | at least 1200 | |||
| DC/DC converter | ||||
| Output voltages, V | at least 15 | |||
| Power, W | at least 1 at least 6 (for models with index M) | |||
| efficiency | at least 80% | |||
| Dynamic characteristics | ||||
| Delay input output t on, µs | no more than 1 | |||
| Residual shutdown delay t off, ms | no more than 0.5 | |||
| Status turn-on delay, µs | no more than 1 | |||
| Recovery time after protection operation, μs | no more than 10 | |||
| not less than 1 (set by capacities Ct, Ct") | ||||
| Response time of the saturation voltage protection circuit when the transistor is turned on tblock, μs | at least 1 | |||
| Threshold voltages | ||||
| min. | type. | Max. | ||
| Threshold of operation of protection for insufficient supply E, V | 10,4 | 11 | 11,7 | |
| The saturation voltage protection circuit of the controlled transistor ensures that the output is turned off and the CT signal is generated at the voltage at the “IK” input, V | 6 | 6,5 | 7 | |
| Insulation | ||||
| Isolation voltage of control signals relative to power signals, V | at least 4000 AC voltage | |||
| DC/DC converter insulation voltage, V | not less than 3000 DC voltage | |||
The proposed drivers allow you to control transistors at a high frequency (up to 100 kHz), which allows you to achieve very high efficiency conversion processes.
Devices of the MD2ХХХ series have a built-in input logic block that allows you to control signals with different values from 3 to 15 V (CMOS) and standard TTL levels, while providing an identical level of transistor gate control signals and forming a delay time for switching the upper and the lower arm of the half-bridge, which makes it possible to ensure the absence of through currents.
Features of the use of drivers on the example of the MD2XXX device
Short review
Driver modules MD215, MD250, MD280, MD215P, MD250P, MD280P are universal control modules designed to switch IGBTs and high-power field-effect transistors.
All types of MD2XXX have mutually compatible contacts and differ only in the level of maximum pulse current.
Higher power MD types - MD250, MD280, MD250P, MD280P are well suited for most modules or several transistors connected in parallel used at high frequencies.
The MD2XXX series driver modules provide a complete solution to control and protection problems for IGBTs and power FETs. In fact, no additional components are required in either the input or the output part.
Action
Driver modules MD215, MD250, MD280, MD215P, MD250P, MD280P for each of the two channels contain:
- an input circuit that provides signal level matching and a protective switching delay;
- electrical isolation between the input circuit and the power (output) part;
- transistor gate drive circuit; on an open transistor;
- circuit for controlling the supply voltage level of the power part of the driver;
- amplifier;
- protection against voltage surges in the output part of the driver;
- electrically isolated voltage source - DC//DC converter (only for modules with P index)
Both driver channels operate independently of each other.
Due to the electrical isolation provided by transformers and optocouplers (subjected to a test voltage of 2650 V AC 50 Hz for 1 min.) between the input circuit and the power section, as well as an extremely high voltage slew rate of 30 kV/µs, driver modules are used in circuits with high potential voltages and large potential jumps occurring between the power section and the control (control) circuit.
The very short delay times of the MD2XXX series drivers make it possible to use them in high-frequency power supplies, high-frequency converters, and resonance converters. Thanks to extremely short delay times, they guarantee trouble-free operation in bridge control.
One of the main functions of the MD2XXX series drivers is a guarantee of reliable protection of controlled power transistors from short circuits and overloads. The emergency state of the transistor is determined by the voltage on the collector of the power transistor in the open state. If a user-defined threshold is exceeded, the power transistor turns off and remains blocked until the end of the active signal level at the control input. After that, the transistor can be turned on again by applying an active level to the control input. This protection concept is widely used for reliable protection of IGBTs.
Functional purpose of conclusions
Pins 14 (BX1 "+"), 13 (BX1 "-")
Pins 13 and 14 are the control inputs of the driver. The control is carried out by applying TTL logic levels to them. Input Vx1 "+" is direct, that is, when a logical 1 is applied to it, the power transistor opens, and when 0 is applied, it closes. Input Vx1 "-" is inverse, that is, when a logical 1 is applied to it, the power transistor closes, and when 1 is applied, it opens. Usually, Vx1 "-" is connected to the common conductor of the input part of the driver, and it is controlled by the input Vx1 "+". The inverting and non-inverting switching of the driver is shown in Fig. 10.
Table 6 shows the state diagram of one driver channel.
Electrical isolation between the input and output parts of the driver on these pins is carried out using optocouplers. Thanks to their use, the possibility of the impact of transients occurring on the power transistor in the control circuit is excluded.
Table 6. State diagram of one driver channel
| Вх1+ | Bx1– | Transistor gate voltage | Transistor saturation voltage > normal | St | St "+ E pit" | Exit |
| X | X | + | X | X | L | L |
| x | x | x | + | l | H | l |
| l | x | x | x | x | H | l |
| x | H | x | x | x | H | l |
| H | l | - | - | H | H | H |
The input circuit has a built-in protection that prevents both half-bridge power transistors from opening at the same time. If an active control signal is applied to the control inputs of both channels, then the circuit will block, and both power transistors will be closed.
Driver modules should be located as close as possible to the power transistors and connected to them with the shortest possible conductors. Inputs Vx1 "+" and Vx1 "-" can be connected to the control and monitoring circuit with conductors up to 25 cm long.
Moreover, the conductors must run in parallel. In addition, the inputs Ix1 "+" and Ix1 "-" can be connected to the control and monitoring circuit using a twisted pair. The common conductor to the input circuit must always be connected separately for both channels to ensure reliable transmission of control pulses.
Whereas reliable transmission of control pulses occurs in the case of a very long pulse, the complete configuration must be checked in the case of a minimum short control pulse.
Conclusion 12 (ST "+ E pit")
Pin 12 is a status output confirming the presence of power (+18 V) on the output (power) part of the driver. It is assembled according to the open collector scheme. During normal operation of the driver (the presence of power and its sufficient level), the status output is connected to the common output of the control circuit using an open transistor. If this status output is connected according to the scheme shown in Fig. 11, then an emergency situation will correspond to a high voltage level on it (+5 V). Normal operation of the driver will correspond to a low voltage level on this status pin. The typical value of the current flowing through the status pin corresponds to 10 mA, therefore, the value of the resistor R is calculated by the formula R \u003d U / 0.01,
where U is the supply voltage. When the supply voltage drops below 12 V, the power transistor turns off and the driver is blocked.
Conclusion 11 (Sz)
An additional capacitor is connected to pin 11, which increases the delay time between the input and output pulses ton on the driver. By default (without an additional capacitor), this time is exactly 1 µs, due to which the driver does not respond to pulses shorter than 1 µs (protection against impulse noise). The main purpose of this delay is to eliminate the occurrence of through currents that occur in half-bridges. Through currents cause heating of power transistors, operation of emergency protection, increase the current consumption, and worsen the efficiency of the circuit. Due to the introduction of this delay by both channels of the driver loaded on the half-bridge, it is possible to control one signal in the form of a meander.
For example, the 2MBI 150 module has a turn-off delay of 3 µs, therefore, in order to exclude the occurrence of through currents in the module when the channels are jointly controlled, it is necessary to put an additional capacitance of at least 1200 pF on both channels.
To reduce the influence of ambient temperature on the delay time, it is necessary to choose capacitors with low TKE.
Pin 10 (ST)
Pin 10 is the status output of the failure on the power transistor of the first channel. A high logical level at the output corresponds to the normal operation of the driver, and a low level - an accident. An accident occurs when the saturation voltage on the power transistor exceeds the threshold level. The maximum current flowing through the output is 8 mA.
Pin 9 (BLOCK)
Pin 6 is the control input of the driver. When a logical unit is applied to it, the driver is blocked and a blocking voltage is applied to the power transistors. The blocking input is common to both channels. For normal operation of the driver, a logical zero must be applied to this input.
Pin 8 is not used.
Pins 7 (+5 V) and 6 (common)
Pins 6 and 7 are inputs for connecting power to the driver. Power is supplied from a source with a power of 8 W and an output voltage of 5 ± 0.5 V. Power must be connected to the driver with short conductors (to reduce losses and increase noise immunity). If the connecting conductors are longer than 25 cm, it is necessary to place noise-suppressing capacitances between them as close as possible to the driver (ceramic capacitor with a capacity of 0.1 μF).
Pin 15 (IR)
Pin 15 (measuring collector) is connected to the collector of the power transistor. Through it, the voltage on the open transistor is monitored. In the event of a short circuit or overload, the voltage across the open transistor rises sharply. When the threshold value of the voltage on the transistor collector is exceeded, the power transistor is blocked and the MT alarm status is triggered. Timing diagrams of the processes occurring in the driver when the protection is triggered are shown in Fig. 7. The protection threshold can be reduced by connecting series-connected diodes, and the threshold value of the saturation voltage U sat. por.=7 –n U pr.VD, where n is the number of diodes, U pr.VD is the voltage drop across the open diode. If the power transistor is powered from a source of 1700 V, it is necessary to install an additional diode with a breakdown voltage of at least 1000 V. The diode cathode is connected to the collector of the power transistor. The protection response time can be adjusted using the output 16-IK1.
Conclusion 16 (IK1)
Pin 16 (measuring collector), unlike pin 15, does not have a built-in diode and limiting resistor. It is necessary to connect a capacitor, which determines the response time of the protection by the saturation voltage on an open transistor. This delay is necessary in order to exclude the effect of interference on the circuit. Due to the connection of the capacitor, the protection response time increases in proportion to the blocking capacitance t =4 С U us. por., where C is the capacitance of the capacitor, pF. This time is added to the driver's internal delay time t off(10%)=3 µs. By default, the driver has a capacitance C = 100 pF, therefore, the protection response delay is t = 4 100 6.3 + t off (10%) = 5.5 µs. If necessary, this time can be increased by connecting a capacitance between pin 16 and the common power supply wire of the power unit.
Pins 17 (out.2) and 18 (out.1)
Pins 17 and 18 are driver outputs. They are designed to connect power transistors and adjust their turn-on time. At pin 17 (out.2), a positive potential (+18 V) is supplied to the gate of the controlled module, and through pin 18 (out. 1), a negative potential (-5 V) is supplied. If it is necessary to provide steep control edges (of the order of 1 µs) and not very high load power (two 2MBI 150 modules connected in parallel), these outputs can be directly connected to the control outputs of the modules. If it is necessary to tighten the fronts or limit the control current (in case of a heavy load), then the modules must be connected to pins 17 and 18 through limiting resistors.
If the saturation voltage exceeds the threshold level, a protective smooth decrease in the voltage at the gate of the control transistor occurs. The time to reduce the voltage at the gate of the transistor to the level of 90%t off (90%)=0.5 µs, to the level of 10%t off(10%)=3 µs. A smooth decrease in the output voltage is necessary in order to exclude the possibility of a voltage surge.
Pins 19 (-E pit), 20 (Common) and 21 (+E pit)
Pins 19, 20 and 21 are the power outputs of the power part of the driver. These pins are supplied with voltage from the DC/DC converter of the driver. In the case of using drivers such as MD215, MD250, MD280 without built-in DC / DC converters, external power supplies are connected here: 19th output -5 V, 20th output - common, 21 output +18 V for current up to 0.2 A.
Calculation and selection of a driver
The initial data for the calculation is the input capacitance of the module C in or equivalent charge Q in, the input resistance of the module R in, the voltage swing at the module input. U = 30 V (given in the reference information for the module), the maximum operating frequency at which the module operates f max.
It is necessary to find the pulsed current flowing through the control input of the module Imax, the maximum power of the DC/DC converter P.
Figure 16 shows the equivalent circuit of the module input, which consists of a gate capacitance and a terminating resistor.
If the charge Q in is specified in the initial data, then it is necessary to recalculate it into an equivalent input capacitance C in =Q in /D U.
The reactive power allocated to the input capacitance of the module is calculated by the formula Рс =f Qin D U. The total power of the DC/DC converter of the driver Р is the sum of the power consumed by the output stage of the driver Рout and the reactive power allocated to the input capacitance of the module Рс: P \u003d P out + Rs.
The operating frequency and voltage swing at the module input were taken as maximum during calculations, therefore, the maximum possible DC/DC converter power during normal driver operation was obtained.
Knowing the resistance of the limiting resistor R, you can find the pulsed current flowing through the driver: I max \u003d D U / R.
Based on the calculation results, it is possible to select the most optimal driver required to control the power module.
Powerful MOSFETs are good for everyone, except for one small nuance - it is often impossible to connect them directly to the microcontroller pins.
This is, firstly, due to the fact that the allowable currents for microcontroller outputs rarely exceed 20 mA, and for very fast switching MOSFETs (with good edges), when you need to charge or discharge the gate very quickly (which always has some capacitance) , we need currents an order of magnitude larger.
And, secondly, the power supply of the controller is usually 3 or 5 Volts, which, in principle, allows you to directly control only a small class of field workers (which are called the logic level - with a logical control level). And given that usually the power supply of the controller and the power supply of the rest of the circuit has a common negative wire, this class is reduced exclusively to N-channel "logic level" field devices.
One of the ways out, in this situation, is the use of special microcircuits - drivers, which are just designed to draw large currents through the gates of the field workers. However, this option is not without its drawbacks. Firstly, drivers are not always available in stores, and secondly, they are quite expensive.
In this regard, the idea arose to make a simple, budget-friendly loose-pack driver that could be used to control both N-channel and P-channel field devices in any low-voltage circuits, say, up to 20 volts. Well, fortunately, I, like real radio junk, in bulk of all electronic junk, so after a series of experiments, this scheme was born:
- R 1 \u003d 2.2 kOhm, R 2 \u003d 100 Ohm, R 3 \u003d 1.5 kOhm, R 4 \u003d 47 Ohm
- D 1 - diode 1N4148 (glass barrel)
- T 1, T 2, T 3 - transistors KST2222A (SOT-23, marking 1P)
- T 4 - transistor BC807 (SOT-23, marking 5C)
The capacitance between Vcc and Out symbolizes the connection of a P-channel field device, the capacitance between Out and Gnd symbolizes the connection of an N-channel field device (the capacitance of the gates of these field devices).
The dotted line divides the circuit into two stages (I and II). In this case, the first stage works as a power amplifier, and the second stage - as a current amplifier. The operation of the circuit is described in detail below.
So. If a high signal level appears at the input In, then the transistor T1 opens, the transistor T2 closes (since the potential at its base drops below the potential at the emitter). As a result, the transistor T3 closes, and the transistor T4 opens and through it the capacitance of the gate of the connected field device is recharged. (The base current of transistor T4 flows along the path E T4 -> B T4 -> D1-> T1-> R2-> Gnd).
If a low signal level appears at the input In, then everything happens the other way around - the transistor T1 closes, as a result of which the potential of the base of the transistor T2 rises and it opens. This, in turn, causes transistor T3 to open and transistor T4 to close. Recharging the capacitance of the gate of the connected field worker occurs through the open transistor T3. (The base current of transistor T3 flows along the path Vcc->T2->R4->B T3 ->E T3).
That's basically the whole description, but some points, probably, require additional explanation.
First, what is the purpose of transistor T2 and diode D1 in the first stage? Everything is very simple here. It was not in vain that I wrote above the paths of the base currents of the output transistors for different states of the circuit. Look at them again and imagine what would happen if there were no transistor T2 with a strapping. Transistor T4 would be triggered in this case by a large current (meaning the base current of the transistor) flowing from the output Out through open T1 and R2, and transistor T3 would be triggered by a small current flowing through resistor R3. This would result in a very long rising edge of the output pulses.
And secondly, many will probably be interested in why resistors R2 and R4 are needed. I stuck them in order to at least slightly limit the peak current through the bases of the output transistors, as well as to finally trim the leading and trailing edges of the pulses.
The assembled device looks like this:
The driver is wired for smd components, and in such a way that it can be easily connected to the main board of the device (in a vertical position). That is, on the main board, we can have a half-bridge, or something else, and already in this board it remains only to vertically insert the driver boards in the right places.
The wiring has some peculiarities. To radically reduce the size of the board, I had to "slightly wrong" to make the wiring of the transistor T4. Before soldering it to the board, you need to turn it face down (marking) and bend the legs in the opposite direction (to the board).
As you can see, the front durations are practically independent of the supply voltage level and are slightly more than 100 ns. In my opinion, pretty good for such a budget design.
